Dynamic power clamp for rfid power control

ABSTRACT

A clamp circuit for an RFID tag includes a power supply node, a dynamic clamp coupled between the power supply node and ground, and an active clamp coupled between the power supply and ground, having a shunt combined effect for providing a clamped power supply node VDDR voltage. The dynamic clamp includes a capacitor divider circuit, a resistor coupled to the capacitor divider circuit, and an N-channel transistor coupled to the capacitor divider circuit. The active clamp includes a differential amplifier having a first input coupled to a resistor divider, a second input for receiving a reference voltage, and an output coupled to a P-channel transistor for the clamped VDDR voltage.

CROSS REFERENCE TO RELATED PATENT APPLICATIONS

The present invention claims priority from U.S. Provisional PatentApplication Ser. No. 61/495,641 filed Jun. 10, 2011, and is related toUnited States patent application Serial Nos. [RAM 608] entitledGENERATION OF VOLTAGE SUPPLY FOR LOW POWER DIGITAL CIRCUIT OPERATION,[RAM 610] entitled POWER-ON SEQUENCING FOR AN RFID TAG, [RAM 611]entitled ANALOG DELAY CELLS FOR THE POWER SUPPLY OF AN RFID TAG, [RAM612] entitled BANDGAP READY CIRCUIT, [RAM 613] entitled DYNAMICADJUSTING RFID DEMODULATION CIRCUIT, and [RAM 614] entitled SHUNTREGULATOR CIRCUIT HAVING A SPLIT OUTPUT, the disclosures of which areherein specifically incorporated by this reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates, in general, to the field of radio frequencyidentification (RFID) tags and systems. More particularly, the inventionrelates to numerous circuit improvements for the RFID tag for optimizingperformance.

2. Discussion of the Related Art

As is well known in the art, a basic RFID system includes threecomponents: an antenna or coil; a transceiver with decoder, i.e., RFIDreader; and a transponder, i.e., RFID tag, programmed with uniqueinformation.

RFID tags are categorized as either active or passive. Active RFID tagsare powered by an internal battery and are typically read/write, i.e.,tag data can be rewritten and/or modified. Passive RFID tags operatewithout a separate external power source and obtain operating powergenerated from the reader.

An example of a typical passive RFID tag is shown in FIG. 1. Tag 100includes an antenna 102 that is coupled to an analog front end circuit104, which is in communication with a digital and memory circuit 106through receive (RX) and transmit (TX) paths. Most passive RFID tagstoday use some sort of electrically erasable programmable read-onlymemory (EEPROM) such as flash memory.

While EEPROM memory has served in passive RFID tag applications to date,the demands for greater data throughput into and out of the RFID areincreasing. This can be seen for example in factory environments, and incollecting highway tolls. The EEPROM based passive RFID tags are slowand may not be suited for the higher throughput applications.Alternative, faster memory technologies such as FRAM (“FerroelectricRandom Access Memory”) exist that are ideally suited for these newhigher speed RFID applications. However, the RFID environment isextremely challenging for FRAM based integrated circuits, not only forthe normal challenges such as the variation in process corners,temperature, and the constraints of low power operation but also forintermittent contact with the RFID reader leading to interruptions withthe available power supply on the RFID tag.

What is desired, therefore, are circuit improvements for an RFID tagthat will provide robust operation in a challenging RFID environmentwhile exploiting the advantages of FRAM memory.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to a clamping circuit foran RFID application that substantially obviates one or more of theproblems due to limitations and disadvantages of the related art.

According to the present invention, a clamp circuit for an RFID tagincludes a power supply node, a dynamic clamp coupled between the powersupply node and ground, and an active clamp coupled between the supplynode and ground, the joint action of dynamic and active clamps providinga well behaved supply node. The dynamic clamp includes a capacitordivider circuit, a resistor coupled to the capacitor divider circuit,and an N-channel transistor coupled to the capacitor divider circuit.The active clamp includes a differential amplifier having a first inputcoupled to a resistor divider, a second input for receiving a referencevoltage, and an output coupled to a P-channel transistor for providingthe regulated VDDR voltage.

A corresponding clamping method for an RFID tag according to the presentinvention includes providing a power supply voltage from an RFIDrectifier having an overshoot in an unclamped condition, and clippingexcess energy harvested by the RFID rectifier during an overshoot timeperiod to prevent the overshoot and to prevent overdriving subsequentRFID circuitry.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory and areintended to provide further explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this specification, illustrate embodiments of the invention andtogether with the description serve to explain the principles of theinvention.

In the drawings:

FIG. 1 is a block diagram of a prior art EEPROM based passive RFID tag;

FIG. 2A is an overall block diagram of a FRAM memory based passive RFIDtag according to the present invention;

FIG. 2B is a more detailed block diagram of a first portion of the FRAMmemory based passive RFID tag referred to in FIG. 2A;

FIG. 2C is a more detailed block diagram of a second portion of the FRAMmemory based passive RFID tag referred to in FIG. 2B;

FIG. 3 is a schematic diagram of the low power voltage regulator and abuffer stage according to the present invention;

FIG. 4 is a schematic diagram of the low power voltage regulator and anemitter follower stage according to the present invention;

FIG. 5 is a graph of VDD_REG vs I_(Load), showing the operation of thelow power voltage regulator according to the present invention;

FIG. 6 is a schematic diagram of a transistor identifying voltage andcurrents used in the graph of FIG. 7;

FIG. 7 is a graph of the sub-threshold operation for a transistor suchas the ones used in the low power voltage regulator according to thepresent invention;

FIG. 8 is a detailed transistor-level schematic of an embodiment of thelow power voltage regulator according to the present invention;

FIG. 9 is a schematic diagram of a clamping circuit for an RFID tagaccording to the present invention;

FIG. 10 is a timing diagram of an overshoot transient in the VDDR powersupply for an RFID due to a slow clamping circuit according to the priorart;

FIG. 11 is a timing diagram of the VDDR response to a fast RF rise withboth active and dynamic clamping according to the present invention;

FIG. 12 is a schematic diagram of a power-on sequencing circuit for anRFID tag according to the present invention;

FIG. 13 is a timing diagram associated with the circuit of FIG. 12;

FIG. 14 is a schematic diagram of a power-on reset circuit according tothe present invention;

FIG. 15 is a timing diagram associated with the circuit of FIG. 14;

FIG. 16 is a block diagram of a state machine according to the presentinvention;

FIG. 17 is a timing diagram associated with the state machine of FIG.16;

FIG. 18 is a timing diagram associated with the power-on reset circuitof FIG. 14;

FIG. 19 is a schematic diagram of a first delay circuit embodiment foruse with a power supply of an RFID tag according to the presentinvention;

FIG. 20 is a timing diagram associated with the delay circuit of FIG.19;

FIG. 21 is a schematic diagram of a second delay circuit embodiment foruse with a power supply of an RFID tag according to the presentinvention;

FIG. 22 is a timing diagram associated with the delay circuit of FIG.21;

FIG. 23 is a schematic diagram of a third delay circuit embodiment foruse with a power supply of an RFID tag according to the presentinvention;

FIG. 24 is a timing diagram associated with the delay circuit of FIG.23;

FIG. 25 is a schematic diagram of a cascaded delay circuit embodimentaccording to the present invention;

FIG. 26 is a timing diagram associated with the cascaded delay circuitof FIG. 26;

FIG. 27 is a schematic diagram of a bandgap ready circuit according tothe present invention;

FIG. 28 is a schematic diagram of a logic circuit for use with thebandgap ready circuit of FIG. 27;

FIG. 29 is a timing diagram associated with the bandgap ready circuit ofthe present invention;

FIG. 30 is a schematic diagram of a prior art demodulation circuit.

FIG. 31 is a timing diagram associated with the prior art demodulationcircuit of FIG. 30;

FIG. 32 is a timing diagram associated with the prior art demodulationcircuit of FIG. 30;

FIG. 33 is a schematic diagram of a dynamic adjusting RFID demodulationcircuit according to the present invention;

FIGS. 34-36 are timing diagrams associated with the dynamic adjustingRFID demodulation circuit according to the present invention;

FIGS. 37A and 37B are schematic diagrams of a prior art shunt regulator;

FIG. 38 is a schematic diagram of a shunt regulator driven from arectifier circuit output having a split output according to the presentinvention; and

FIGS. 39 and 40 are timing diagrams associated with the shunt regulatorof FIG. 38.

DETAILED DESCRIPTION OF THE ILLUSTRATED EMBODIMENTS

Referring now to FIG. 2A, a passive RFID tag 200 according to theinvention includes an antenna 202, an analog front end 204, and adigital portion 206 that includes digital control circuitry and FRAMmemory and communicates with the analog front end 204 using the RX andTX paths.

Referring to FIG. 2B, a more detailed block diagram of a first portion208 of the RFID tag 200 includes a rectifier 210, a split output 212including two diodes described in further detail below, an active clamp214, and a dynamic clamp 218 coupled to the VDDR power supply, which isalso described in further detail below. The rectifier output is alsocoupled to a demodulator 216 for providing a digital output, which isalso described in further detail below.

Referring to FIG. 2C, a more detailed block diagram of a second portion220 of the RFID tag 200 includes a slew filter 224 and a bandgap circuit222 both coupled to the VDDR power supply, and both described in furtherdetail below. The output of the slew filter is coupled to a VDDMregulator 226 for providing a VDDM supply voltage. In turn, a VDDDregulator 228 is coupled to the VDDM supply voltage, as well as delaycircuits 232 and 234. The purpose and nature of these circuits isdescribed in further detail below. A VDDMPOK circuit 238 receives theVDDM, VBG, and DLY2 signals and provides a VDDMPOK signal. A delaycircuit 240 receives the VDDMPOK signal and provides a GEN2POK signal. Areset circuit 242 receives the GEN2POK and VDDMPOK signals to provide aRESET signal. Finally, circuit 236 receives the VDDMPOK, GEN2POK, andPORBTHRESH signals to provide a VDDD_PORB signal. The nature of all ofthese further signals and circuits is explained in further detail below.Circuit 230 monitors the VDDD threshold.

Referring now to FIG. 3 a low power voltage regulator 300 includes anoutput node for providing a regulated output voltage (VDD_REG), a firstdiode-connected transistor Q1 of a first polarity type (P-channel) inseries with a second diode-connected transistor Q2 of a second polaritytype (N-channel) coupled between the output node and ground. A biascurrent I_(BIAS) has a value for biasing the first and seconddiode-connected transistors in a sub-threshold mode of operation as isexplained in further detail below. The I_(BIAS) current can be generatedusing a bandgap circuit or other bias current circuits as are known inthe art. A buffer amplifier 302 is coupled to the output node to providea low impedance regulated output voltage.

Referring now to FIG. 4 a low power voltage regulator 400 includes anoutput node for providing a regulated output voltage (VDD_REG), and afirst diode-connected transistor Q1 of a first polarity type (P-channel)in series with a second diode-connected transistor Q2 of a secondpolarity type (N-channel) coupled between the output node and ground. Adiode D3 is provided to compensate for the voltage drop seen in Q3 tothe regulated voltage VDD_REG. A bias current I_(BIAS) has a value forbiasing the first and second diode-connected transistors in asub-threshold mode (or other mode) of operation as is explained infurther detail below, as well as diode D3. An emitter follower stagetransistor Q3 is coupled to the output node (through diode D3) toprovide a low impedance regulated output voltage.

Referring now to FIG. 5, it is important to note that the total voltageacross transistors Q1 and Q2 is not a constant voltage, even though thebias current, labeled I_(BIAS), may be. Thus, the circuit of FIG. 4 isnot a constant voltage generator. At lower temperatures a higher totalvoltage V₂ results given a constant bias current. At higher temperaturesa lower total voltage V₁ results given the same constant bias current.Thus, the total voltage decreases with increased temperature change. Thevoltage VDD_REG is a temperature compensated voltage, which helps toprovide stable circuit performance. A higher VDD_REG voltage is providedto compensate for relatively slower transistors, which occurs at lowertemperatures. A lower VDD_REG voltage is provided to compensate forrelatively faster transistors, which occurs at higher temperatures. Thelevel of I_(BIAS) also sets the operating mode of Q1 and Q2 assub-threshold or higher power mode.

Referring now to FIG. 6, a transistor is shown identifying thedrain-source current (I_(DS)), drain-source voltage (V_(DS)), and thegate-source voltage (V_(GS)) thereof.

Referring now to FIG. 7, a graph is shown plotting the drain-sourcecurrent of a transistor in response to the gate-source voltage. Athreshold voltage V_(THRESHOLD) is shown. Above the threshold voltage,the transistor operates in the “Square Law” mode of operation. Below thethreshold voltage, the transistor operates in an exponential mode ofoperation, with the leakage current of the device forming a currentfloor level. In the circuit of FIG. 4, the bias current throughtransistors Q1 and Q2, as well as serially connected diode D3, isconstrained to operate those transistors in the sub-threshold mode ofoperation for very low power operation. This insures that the circuit ofFIG. 4 operates a low power mode, which is of critical importance in anRFID tag, as the available energy for circuit operation is extremelylimited. With larger I_(BIAS) a higher power, higher performance circuitoperation is obtained.

Referring now to FIG. 8, a detailed transistor-level schematic of anembodiment of the low power voltage regulator 800 is shown. Low powervoltage regulator 800 includes a first diode-connected transistor Q1 ofa first polarity type (P-channel) in series with a seconddiode-connected transistor Q2 of a second polarity type (N-channel)coupled between the output node and ground. A bias current generator I₁is used to bias the first and second diode-connected transistors in asub-threshold mode of operation. A third diode-connected transistor (Q3,N-channel) is coupled between the bias current I₁ and the first andsecond diode-connected transistors Q1 and Q2. A buffer amplifier (Q4,Q5, I₂, I₃) is coupled to the third transistor Q3 for providing aregulated output voltage VDD_REG. The buffer amplifier includes an inputtransistor Q4 (N-channel) having its gate forming the input of thebuffer, and its source forming the output of the buffer for providingthe regulated output voltage. A feedback transistor Q5 (P-channel) isalso included having a gate coupled to the drain of the inputtransistor, a source for coupling to a power supply voltage, and a draincoupled to the source of the input transistor Q4. A first buffer biascurrent I₂ is coupled to the drain of the input transistor Q4 and asecond buffer bias current I₃ is coupled to the source of the inputtransistor Q4. The value of the second bias current I₃ is greater thanthe value of the first bias current I₂. Representative values of thebias currents for the voltage regulator 800 in very low power operationare as follows: I₁ is 20 nA, I₂ is 20 nA, and I₃ is 40 nA.

A clamp circuit 900 for an RFID tag is shown in FIG. 9 including a VDDRpower supply node (clamped power supply voltage provided by the RFIDrectifier circuit, not shown in FIG. 9), a dynamic clamp 902 coupledbetween the power supply node and ground, and an active clamp 904 alsocoupled between the power supply and ground, holding the power supplyVDDR at a maximum steady state clamped value when the RF supplied powerexceeds an operating minimum.

The dynamic clamp 900 includes a capacitor divider circuit includingcapacitors C91 and C92 coupled between the VDDR rail and ground. Aresistor R91 is coupled to the capacitor divider circuit at center tapnode 906. An N-channel transistor Q91 has a gate coupled to thecapacitor divider circuit at center tap node 906. The drain oftransistor Q91 is coupled to the VDDR rail and the source is coupled toground.

The active clamp includes a differential amplifier 906 having a firstinput coupled to a resistor divider including resistors R92 and R93 atcenter tap node 908, a second input for receiving a reference voltageVREF, and an output coupled to a P-channel transistor Q92 for providingthe clamped VDDR voltage. The differential amplifier 906 can be anoperational amplifier. The gate of transistor Q92 is coupled to theoutput of differential amplifier 906, the clamped VDDR voltage isprovided at a source of the transistor, and the drain of the transistoris tied to ground. A holding capacitor C93 is attached between VDDR andground. The clamped VDDR is also referred to as ‘VDD’ in the followingsection.

Referring now to FIG. 10, the overshoot transient in the VDD clampedsupply voltage is shown due to the undesirably slow response of a priorart clamping circuit. The power supply voltage waveform 1002 attains adesirably final VDD voltage level only after experiencing a significantovershoot that may adversely affect downstream circuitry.

Referring now to FIG. 11, the prior art VDD voltage waveform 1102including the undesirable overshoot is shown along with the VDD voltagewaveform 1104 clamped with the active and dynamic clamping according tothe present invention. Note that the voltage waveform 1104 has a muchreduced overshoot.

Thus, a clamping method for an RFID tag according to the presentinvention includes providing a power supply voltage from an RFIDrectifier having an overshoot in an unclamped condition, and clippingexcess energy harvested by the RFID rectifier during an overshoot timeperiod to prevent the overshoot and to prevent overdriving subsequentRFID circuitry. This method is provided by using a dynamic clamp incascade with an active clamp. The dynamic clamp includes an NMOStransistor for shunting fast rising initial energy from the RFIDrectifier output, and further includes a leakage path for turning offthe NMOS transistor after the overshoot time period. Stated another way,a clamping method for an RFID tag according to the present inventionincludes removing energy harvested by an RFID rectifier from a fastrising RF field that would generate an overshoot condition for apredetermined initial time period to prevent overshoot and to preventoverdriving subsequent RFID circuitry.

Referring now to FIG. 12, circuit 1200 provides sequencing control foran RFID tag according to the present invention. The voltage at inputnode (1) is the raw power supply voltage VDDR supplied by thedynamically and actively clamped output of the rectifier, Power VDDR inFIG. 2B. The circuit 1200 can be realized in a one-chip or two-chipsolution. Circuit block 1202 is a normal bandgap circuit, coupled to thebandgap ready circuit 1204 at node (2). Bandgap ready circuit 1204 isexplained in further detail below. Circuit block 1206 is a slew filterthat is explained in further detail below. The output of the slew filteris shown as node (3), coupled to filter capacitor 1208. Circuit block1210 is an LDO regulator generating power for the memory circuit block1224, for the digital voltage regulator circuit block 1212. It alsoprovides a divided down signal to circuit block 1214, a comparator,where it is compared to the bandgap voltage to generate signal VDDMPOKexplained further below. The “memory VDDM” voltage means the VDD voltageprovided to the memory. Note FRAM memory block 1224 is coupled to theVDDM line. Circuit block 1212 is a second regulator for providing theregulated VDDD voltage to the digital circuitry on the chip designatedby block 1226 at node (5). Circuit block 1214 is a comparator. Threeinputs are shown, which include a positive input, a negative input, andan enable input. The outputs of comparator 1214 are VDDMPOK, and theinverted VDDMPOK signals. The VDDMPOK signal node is labeled (6A). TheVDDMPOK signal designation means “VDDM Power is OK”. Blocks 1216 and1218 are delay circuits that are explained in further detail below.Circuit block 1220 is also a delay circuit. Delay circuit 1220 can be asimple analog delay using a current source and a capacitor, and is usedto generate the GEN2POK signal at node (6B) as shown. Block 1222 is avoltage monitor, and is described in further detail below. Finally, areset signal generating block 1228 is shown for generating a resetsignal at node 1808.

FIG. 13 is a timing diagram associated with FIG. 12, wherein the nodewaveforms for nodes (1), (2), (3), (4), and (5) are shown. Thecorresponding waveforms are designated 1302, 1304, 1306, 1308, and 1310.One of the most important functions associated with the circuit of FIG.12 is to protect the FRAM memory from any loss of contents by shuttingdown the memory properly. That is, a new memory access is prevented whenthe memory voltage is below a certain value. Memory operation is onlypossible when VDDM is above a first threshold value, and maintains thatvalue. Memory operation is discontinued when VDDM drops below a secondthreshold value.

Referring now to FIG. 14, a voltage monitor circuit is shown including atransistor Q1402, a capacitor 1404, and a digital circuit includinginverters 1406 and 1408, and an OR gate 1410 for generating a PORsignal.

Referring now to FIG. 15, a waveform 1502 is shown that is associatedwith the monitor circuit of FIG. 14.

Referring now to FIG. 16, a state machine 1600 is shown that receivesall of the various timing inputs and generates the VDDD_PORB signal.State machine 1600 operates according to the follow rules:

If PORB_THRESH is low, the output of the state machine is low.

If PORB_THRESH is high, then:

If the other two input signals are high, then the output is high.

If the other two input signals are low, then the output is low.

If only one of the two input signals are high, the output is high.

Referring now to FIG. 17, a timing diagram is shown associated with thestate machine of FIG. 16. In particular, the fall edge delay withrespect to the VDDMPOK signal 1702, and the GEN2POK signal 1704 isshown, which is in the range of about 2 to 6 microseconds.

Referring now to FIG. 18, a timing diagram associated with the overallsequencing circuit of FIG. 12 is shown including the input VDDRAWvoltage, the VDDMPOK signal 1804, the GEN2POK signal 1806, and thedischarge pulse 1808.

Referring now to FIG. 19, a first delay circuit 1900 for use with apower supply in an RFID tag includes a power supply input VDD_IN and apower supply output VDD_OUT. A passive circuit 1902, 1904 is coupledbetween the power supply input VDD_IN and ground. A transistor Q1906 hasa current path coupled between the power supply input VDD_IN and thepower supply output VDD_OUT, and a control node coupled to anintermediate node 1908 of the passive circuit. The passive circuitincludes a capacitor 1902 and a resistor 1904 in series connection. Thecapacitor 1902 is coupled between the power supply input VDD_IN and theintermediate node 1908. The resistor 1904 is coupled between theintermediate node 1908 and ground. Transistor Q1906 is a P-channeltransistor.

Referring now to FIG. 20, the response of the delay circuit 1900 isshown. The VDD_IN typically supplied by a diode rectifier on the RFIDtag and has an overshoot indicated by waveform 2002. The VDD_OUTwaveform 2004 after being processed by first delay circuit 1900 has noovershoot, and is delayed by a predetermined delay time perioddetermined by the time constant of the passive circuit includingcapacitor 1902, resistor 1904.

Referring now to FIG. 21, a second delay circuit 2100 for use with apower supply in an RFID tag includes a power supply input VDD_IN and apower supply output VDD_OUT. A ramp circuit 2102, 2104 is coupledbetween the power supply input VDD_IN and ground. A transistor Q2106 hasa current path coupled between the power supply input VDD_IN and thepower supply output VDD_OUT, and a control node coupled to anintermediate node 2108 of the ramp circuit. The ramp circuit includes acapacitor 2102 and a current source 2104 in series connection. Thecurrent source can be a temperature stabilized current source providedfrom a bandgap circuit if desired. The capacitor 2102 is coupled betweenthe power supply input VDD_IN and the intermediate node 2108. Thecurrent source 2104 is coupled between the intermediate node 2108 andground. Transistor Q2106 is a P-channel transistor.

Referring now to FIG. 22, the response of the second delay circuit 2100is shown. The VDD_IN typically supplied by a diode rectifier on the RFIDtag and has an overshoot indicated by waveform 2202. The VDD_OUTwaveform 2204 after being processed by second delay circuit 2100 has noovershoot, and is delayed by a predetermined delay time perioddetermined by the ramping speed of the ramp circuit including capacitor1902, resistor 1904. The predetermined delay time includes a first delaytime DLY1, which is determined by the current source turning on and asecond delay time DLY2, which is determined by transistor Q2106 turningon. The bandgap waveform 2206 is also shown in FIG. 22.

Referring now to FIG. 23, a third delay circuit 2300 for use with apower supply in an RFID tag includes a power supply input VDD_IN and apower supply output VDD_OUT. A current mirror circuit Q2302, Q2304 hasan input coupled to a current source 2306, an output coupled to thepower supply output VDD_OUT, and a power node coupled to the powersupply input VDD_IN. A capacitor 2308 (C_(LARGE)) is coupled between thepower supply output VDD_OUT and ground. The current mirror circuitcomprises a simple two-transistor current mirror with a P-channel MOSinput transistor Q2302 and a P-channel MOS mirror transistor Q2304.Other more complicated current mirror circuits as are known in the artcan also be used. Current source 2306 can be a temperature stabilizedcurrent source from bandgap circuit if desired.

Referring now to FIG. 24, the response of the third delay circuit 2300is shown. The VDD_IN typically supplied by a diode rectifier on the RFIDtag and has an overshoot indicated by waveform 2402. The VDD_OUTwaveform 2404 after being processed by first delay circuit 1900 has noovershoot, and is delayed by a predetermined delay time perioddetermined by I_(REF) turning on. The slew rate of the output waveform2404 is defined by the values of the current source 2306 and thecapacitor 2308 until a stable final output voltage value is reached.

A cascaded delay circuit providing power to a regulator 2600 for an RFIDtag is shown in FIG. 25 including a power supply input VDD_IN and apower supply output VDD_OUT and two delay circuits 2602, 2604 in cascadeconnection between the power supply input and the power supply output.Different combinations of cells and numbers can be used. For example,delay cell 2602 can be the delay cell 1900 shown in FIG. 19 or the delaycell 2100 shown in FIG. 21. The delay cell 2604 can be the slew filter2300 shown in FIG. 23. The regulator 2606 can be any known voltageregulator such as an LDO, shunt, or source follower regulator.

Delay cell 2602 can comprise delay circuit 1900 or delay circuit 2100 asdiscussed above coupled between a local power supply input (VDD_IN) anda local power supply output 2610.

Slew filter 2604 can comprise delay circuit 2300 as discussed abovecoupled between a local power supply input 2610 and a local power supplyoutput 2612.

Voltage regulator 2606 can comprise any known voltage regulator asdiscussed above coupled between a local power supply input 2612 and alocal power supply output VDD_OUT.

Referring now to FIG. 26, a number of response waveforms are shown inthe timing diagram associated with circuit 2600 in FIG. 25. The VDD_INwaveform 2702 is shown having an overshoot. The output of the firstdelay cell 2706 is shown slightly delayed and having no overshoot. Theoutput of the slew filter 2704 is shown still further delayed and havinga slew-controlled output up to a final stable voltage output. Thisoutput is still further delayed and regulated by the voltage regulatoras shown in waveform 2708. The total delay 2710 is shown between theonset of the VDD_IN waveform and the start of the VDD_OUT regulatedoutput voltage.

The purpose of the various single and cascaded power supply delaycircuits is to provide a control mechanism for turning on circuit blocksand functions inside of an RFID tag only when it can be assured that astable power supply voltage can be provided. It will be apparent tothose skilled in the art that the cascaded delay circuit 2600 can bedesigned with other arrangements of delay cells while still providingthe desirable stable voltage function.

Referring to FIG. 27, a circuit for detecting the safe voltage operationfor a chip such as found in an RFID tag, it is necessary to first detectwhen the system reference level, derived from a bandgap voltagegenerator, is sufficiently stable so that signals generated from thisreference will be in a range near steady state operation. A regulatorthat has a bandgap reference generates a voltage proportional to thereference. If the reference is not fully settled, the regulator outputis not in its design range. The bandgap circuit operates by feedbackcontrol to maintain operation at a crossing point of two node voltages,nodes 1 and 2. According to the present invention, a third branch thatcrosses one of the nodes during the turn-on transient and is at a highersteady state voltage, at a lower than final operational voltage is usedto generate a voltage that is compared with one of the bandgap referencevoltages, node 1, to create part of a bandgap ready logic signal.However, when addressing input transients, slow and erratic powersupplies, an additional problem was identified. The branch transientsare not well controlled and an erroneous valid operation was predicted.To fix this secondary problem, an additional monitoring circuit wasadded to detect saturation operation of the core branch of currents inthe bandgap voltage generator that would be combined with the crossinginformation logic signal to more reliably predict when the bandgapreference cell was close to steady state operation. Once the band gapreference cell is at steady state operation valid comparisons ofregulator outputs for signaling to control circuits that the properstate of the power supply has been reached.

In the bandgap ready circuit 2800 of FIG. 27, levels that are monitoredfor control of operations only at a defined range rely on having areference that is at or close to the end of its turn-on transient. Oncea reference valid monitor for the bandgap reference cell is generated,false release of operation of the RFID chip at voltage levels that aretoo low for reliable operation is substantially obviated.

Bandgap circuits are well known in the art. It is also well known thatvoltage regulation such as is required in a chip in an RFID tag requiresa stable bandgap reference voltage. During power up, monitor circuitsused in the power-up sequence will undesirably glitch if the referenceis not yet stable and may improperly release circuit functions whensupplies are out of operational range. Prior art uses timing delays toallow the bandgap reference circuit to settle before indicating that astable operating voltage has been reached. While the approach accordingto the present invention is effective for assuring that a properoperating voltage has been reached, and then other circuit function canbegin, it is process sensitive and should be tuned to ensure optimumperformance.

Referring to FIG. 27, a ‘bandgap ready’ circuit 2800 includes a bandgapcircuit for providing a bandgap voltage including diodes D2820, D2822,resistor R1 designated 2832, N-channel transistors Q2810 and Q2812, andP-channel transistors Q2804 and Q2806. Transistors Q2810 and Q2812 forman N-channel current mirror. Transistors Q2804 and Q2806 form aP-channel current mirror. Other bandgap designs would have equivalentmonitoring nodes from this example design. P-channel transistor Q2808 ismirrored from the P-channel current mirror. The drain current fromtransistor Q2808 is used to generate the V_(BGAP) voltage across dioderesistor R2 designated 2834 and diode D2824. A capacitor C2826 iscoupled to the bandgap output voltage node. A first comparator 2828 isused for monitoring first and second voltages in the bandgap circuit andfor providing a first logic signal at node 2814. A first input iscoupled to the source of transistor Q2810 and a second input is coupledto the drain of P-channel transistor Q2802 in a replica branch. Thecurrent through transistor Q2802 generates a slightly larger andtracking voltage to node 2 in the bandgap core, across resistor R3designated 2830 and diode D2818. A second comparator 2826 is used formonitoring third and fourth voltages in the bandgap circuit and forproviding a second logic signal at node 2816. A first input is coupledto the drain of transistor Q2806 and a second input is coupled to thedrain of transistor Q2808, which is also coupled to resistor R2830, asshown.

In circuit 2800 shown in FIG. 27, the relative sizes of the diodes forproper generation of the bandgap voltage are given as follows: D2818,m=M; D2820, m=1; D2822, m=N; and D2824, m=1.

Referring now to FIG. 28, an AND logic circuit 2900 for combining thefirst and second logic signals at nodes 2814 and 2816 is used to providea bandgap ready logic signal BGOK.

Referring now to FIG. 29( a) a graph of the voltages at the source oftransistor Q2810 and drain of transistor Q2802 are shown, designatednodes (1) and (3006) in FIG. 27. The crossover point of these twovoltages is used to generate the first logic signal at node 2814. Thecomparison is not done directly on core nodes, but rather through thereplica branch provided by transistor Q2802, resistor R3 designated2830, and diode D2818 as discussed above. In FIG. 29( b) the VDD supplyvoltage is shown crossing a minimum allowable operational threshold withrespect to time. Waveform 3002 represents the voltage at node (3),waveform 3004 represents the voltage at node (1), waveform 3006represents the voltage at the top of resistor R3 at node (2), andvoltage difference 3008 represents a guaranteed crossing voltagedifference due to steady state core current I multiplied by R3.

Referring now to FIG. 30, a prior art demodulation circuit 3100 is shownincluding an RF input at node 3114, an input diode 3102 coupled betweeninput node 3114 and node 3116. A resistor 3104 in parallel withcapacitor 3106 is coupled to node 3116. Diode 3102, resistor 3104, andcapacitor 3106 form an envelope detector as is known in the art. Thesignal at node 3116 is filtered by a low pass filter including resistor3108 and capacitor 3110. The output of the low pass filter at node 3118,and the envelope signal at node 3116 are received by a comparator 3112to provide a data output digital signal at output node 3120.

Referring now to FIG. 31, a timing diagram is shown including the RFwaveform 3114 and the envelope waveform 3116. The timing diagram of FIG.32 shows the envelope waveform 3116, as well as the output waveformshowing the desirable average value and the undesirable ripple that isproduced due to the averaging circuit 3100.

Prior art RFID demodulation circuits such as those described above maynot provide proper operation over all input power levels due to largeinput signals at high power. Averaging schemes to correct this problemare problematic because they are data rate dependent, leading to avariation in pulse widths during the averaging transient. To correctthis problem, a fixed reference was added where the power levelreduction in operating margin was detected. An additional signaldependent current was added to the reference voltage so that higherpower levels generated their own higher level reference as is explainedin further detail below.

Transient pulse width changes with averaging circuits corrupt datadetection. With existing averaging circuits a single filter timeconstant is not sufficient for both low and high data rates. At highdata rates the ripple signal on the average is low, but there is a longtransient during which the duty cycle changes. At low data rates theripple signal is large and can cross the detection threshold dependingon input power level. Both of these extremes can see errors in datadetection during startup conditions.

Referring now to FIG. 33, a dynamic adjusting RFID demodulation circuit3400 is shown according to the present invention. Dynamic adjusting RFIDdemodulator circuit 3400 includes an envelope detector 3402, 3404, 3406having an input for receiving a modulated RF signal at node 3422, afixed reference 3412, Q3414 coupled to the input of an RC filter 3416,3418 and a comparator 3420 having a first input coupled to the output ofthe envelope detector, a second input coupled to an output of the RCfilter, and an output for providing a data output signal at node 3430.The envelope detector includes an input diode 3402, parallel resistor3404 and capacitor 3406, as well as an output diode 3408 and an outputresistor 3410. The fixed reference includes a current source 3412, whichcan be a thermally compensated current source derived from a bandgapcircuit, and a diode-connected N-channel transistor Q3414. Other fixedreferences can be used. The RC filter includes a resistor 3416 and acapacitor 3418. The first input of the comparator 3424 is a positiveinput and the second input is a negative input in a preferredembodiment.

Demodulator circuit 3400 of the present invention thus includes a fixedreference generated by a current source (from a bandgap circuit ifdesired) and a diode-connected MOS transistor and an RC filter plusanother energy path activated at higher power levels that inject currentinto the output of the RC filter at the input to the comparator. Thethreshold of the comparator is thus augmented proportionally to theinput power so that high power and low power RF input signals areequally discriminated.

Referring now to FIGS. 34-36, various circuit waveforms are shown thatillustrate the dynamic threshold of circuit 3400 responding to differentinput levels. In FIG. 34, two RF envelopes are shown. A first RFenvelope 3424A is shown at low power levels. A second RF envelope 3424Bis shown at relatively higher power levels. The threshold voltage forcomparator 3420 at low power levels is the same at both nodes 3426 and3428. However, as is shown in FIG. 35, the voltage levels at nodes 3426and 3428 are different at higher power levels. That is, the voltage atnode 3428 is greater than that at node 3426 at higher power levels.Finally, in FIG. 36, the voltage at node 3428 is shown with respect toincreasing RF input power levels. The voltage at node 3428 begins at afirst level and increases as the input power is increased.

Referring now to FIG. 37A, a schematic diagram of a prior art shuntvoltage regulator 3800 is shown in the context of an RFID tagapplication. A rectifier 3802 receives an RF input signal, which isrectified to provide a supply voltage, which is the same voltage nodethat is regulated, V_(REG), as is known for a shunt regulator. Aresistor divider circuit and comparator 3804 are coupled between V_(REG)and ground. The resistor divider circuit and comparator receive an inputreference voltage and provide a control voltage at node 3812 for adischarge device. A P-channel discharge device shown here for example,transistor Q3806 has a source coupled to the V_(REG) node, a gate forreceiving the control voltage at node 3812, and a drain coupled toground. The V_(REG) is coupled to a large holding capacitor 3808, whichprovides a stable voltage and energy for powering on-chip circuits suchas a FRAM memory circuit, various digital and analog circuits, and I/Ocircuits.

The resistor divider circuit and comparator 3804 are shown in furtherdetail in FIG. 37B, wherein a resistor divider including resistors 3814and 3816 is coupled between the V_(REG) and ground. An amplifier 3818receives the input reference voltage at a first input (negative) and atap voltage of the resistor divider at a second input (positive). Theoutput voltage of the amplifier 3818 provides the control voltage forthe P-channel transistor Q3806 at node 3812 as shown.

The holding capacitor 3808 is effectively the on-chip power supplyvoltage for the rest of the integrated circuit, or circuits in an RFIDtag. The voltage on the capacitor 3808 is from charge harvested from anRF reader. It is important that this charge be conserved and not wastedduring any regulation operations.

Referring now to FIG. 38, a shunt regulator 3900 for an RFID tag chipaccording to the present invention is shown having a split source outputfrom the RF rectifier including a first output 3914 for providing apower delivery path to on-chip circuits 3916 and a second output 3924for providing a discharge-regulation path. As previously discussed, theon-chip circuits can include a FRAM memory circuit, I/O circuits, andother digital and analog circuitry as required for a specificapplication. A large holding capacitor 3912 is coupled between the firstoutput 3914 and ground. The shunt regulator 3900 includes an input node3902 for receiving a power supply voltage from a rectifier output, afirst diode 3904 having an anode coupled to the input node, a seconddiode 3906 having an anode coupled to the input node, a resistor dividercircuit and comparator 3908 coupled between a cathode of the first diodeand ground, a P-channel transistor 3910 having a control terminalcoupled to an output of the resistor divider circuit and comparator atnode 3918, and a current path coupled between a cathode of the seconddiode and ground, wherein the cathode of the first diode forms the firstoutput 3914 and the cathode of the second diode forms the second output3924.

The resistor divider and comparator circuit 3908 are substantially thesame as is shown in FIG. 37B.

Referring now to FIG. 39 a plot 4000 is shown of the unregulated voltagewaveform 4002 and the regulated voltage waveform 4004 with respect toground. The input unregulated voltage 4002, which exceeds a desirableupper value, is shown to be regulated to a constant acceptable uppervalue in the regulated voltage 4004. Note that current I_(CONTROL),shown in FIG. 38 pulls down node 3902, isolating node 3914 in thepresence of excess RF energy.

Referring to FIG. 40, the input voltage from the rectifier at node 3902is plotted in juxtaposition with the output voltage at the first outputnode 3914 with the shunt regulator crossing into regulation. Anovervoltage condition results in activation of Q3910 in FIG. 39 dumpingexcess harvested energy and pulling down node 3902 and isolating 3914and not discharging capacitor 3912. Note that in the case of a voltagedrop-out due to loss of contact with the RF reader, the input voltagealso drops; however, the on-chip power supply voltage at node 3914remains high momentarily, and the extra charge is conserved and can becontinued to be used for powering on-chip circuits. With the separationof the discharge-regulation path from the power delivery path regulationcan be maintained while not removing the charge from the hold capacitor3912 that powers the rest of the circuitry. Separating the outputsaccording to the present invention as described makes the RFID tag moreefficient.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the invention. As would beapparent to those skilled in the art, equivalent embodiments of thepresent invention can be realized in firmware, software, or hardware, orany possible combination thereof. In addition, although representativeblock diagrams are shown for an aid in understanding the invention, theexact boundaries of the blocks may be changed and combined or separatedout as desired for a particular application or implementation. Finally,although FRAM memory is described and claimed, the present invention isalso applicable to any other high speed non-volatile memory technology.Thus, it is intended that the present invention cover the modificationsand variations of this invention provided they come within the scope ofthe appended claims and their equivalents.

1. A clamp circuit for an RFID tag comprising: a power supply node; adynamic clamp coupled between the power supply node and ground; and anactive clamp coupled between the power supply node and ground, whereinthe combined action of the dynamic and active clamps moderate thebehavior of the power supply node to fast rising RF inputs.
 2. The clampcircuit of claim 1 wherein the dynamic clamp comprises a capacitordivider circuit.
 3. The clamp circuit of claim 2 further comprising aresistor coupled to the capacitor divider circuit.
 4. The clamp circuitof claim 2 further comprising a transistor coupled to the capacitordivider circuit.
 5. The clamp circuit of claim 4 wherein the transistorcomprises an N-channel transistor.
 6. The clamp circuit of claim 1wherein the active clamp comprises a differential amplifier having afirst input coupled to a resistor divider, a second input for receivinga reference voltage, and is coupled to a transistor for providing theclamped VDDR voltage.
 7. The clamp circuit of claim 6 wherein thedifferential amplifier comprises an operational amplifier.
 8. The clampcircuit of claim 6 wherein the differential amplifier comprises acomparator.
 9. The clamp circuit of claim 6 wherein the transistorcomprises a P-channel transistor.
 10. The clamp circuit of claim 9wherein the clamped VDDR voltage is provided at a source of theP-channel transistor.
 11. A clamping method for an RFID tag comprising:providing a power supply voltage from an RFID rectifier having anovershoot in an unclamped condition; and clipping excess energyharvested by the RFID rectifier during an overshoot time period toprevent the overshoot and to prevent overdriving subsequent RFIDcircuitry.
 12. The clamping method of claim 11 further comprisingproviding a dynamic clamp in cascade with an active clamp.
 13. Theclamping method of claim 12 wherein the dynamic clamp comprises acapacitive divider function.
 14. The clamping method of claim 12 whereinthe dynamic clamp comprises an NMOS transistor for shunting initialenergy from the RFID rectifier output.
 15. The clamping method of claim14 further comprising a leakage path for turning off the NMOS transistorafter the overshoot time period.
 16. A clamping method for an RFID tagcomprising removing energy harvested by an RFID rectifier for apredetermined initial time period to prevent overshoot and to preventoverdriving subsequent RFID circuitry.
 17. The clamping method of claim16 further comprising providing a dynamic clamp in cascade with anactive clamp.
 18. The clamping method of claim 17 wherein the dynamicclamp comprises a capacitive divider function.
 19. The clamping methodof claim 17 wherein the dynamic clamp comprises an NMOS transistor forshunting initial energy from the RFID rectifier output.
 20. The clampingmethod of claim 19 further comprising a leakage path for turning off theNMOS transistor after the overshoot time period.